1. Field of the Invention
This invention relates generally to microwave integrated filter tuning systems. More specifically, this invention relates to an improved circuit for automatically tuning and controlling loss in voltage-controlled oscillator (VCO) master-slave tuning systems incorporating microwave integrated filters.
2. Description of the Related Art
Interest in building single-chip wireless transceivers has fueled the integration of microwave continuous-time filters, of which automatic tuning systems are crucial parts. These tuning systems correct time constants and compensate for excess phase shift to make the filters immune to parasitics, technology tolerances, and temperature changes. One tuning approach uses a VCO and a frequency control circuit to set the time constant of a tracking filter. This system requires amplitude regulation to reduce distortion caused by the non-linearities of the active devices in the VCO in order to minimize the tuning error. Two major methods of regulation exist: (1) using limiters to limit the amplitude and (2) using a second control circuit, in addition to the frequency control circuit. Because of the non-linearity of the limiters, the first method causes a large frequency tuning error as the VCO's quality factor (Q) decreases. The second control circuit is used in the second method to make Q infinite and the amplitude small, and all elements of the second control circuit are intended to work in the linear region. In microwave integrated filter applications involving GHz-range coupled-resonator bandpass filters, the loss of the VCO must be tuned to zero.
The latter type of tuned-filter system discussed above is illustrated in FIG. 1. This system 10, often called a VCO master-slave tuning system, includes tuned filter 100 (also called an "indirectly" tuned filter), VCO 110, frequency control circuit 120, and Q-control circuit 130. VCO 110 is the "master" and tuned filter 100 is the "slave." The circuit topologies of the master and the slave are very similar, often including transconductance amplifiers, bandpass filters, coupled resonators, or a combination of these or other elements. Tuned filter 100 includes f and Q inputs which control the center frequency (in the case of a bandpass filter) and Q of tuned filter 100. These f and Q inputs are also provided to VCO 110. Frequency control circuit 120 is provided with a reference frequency F.sub.REF and typically includes a phase detector and a low-pass filter. When frequency control circuit 120 is combined in a loop with VCO 110, a phase-locked loop results, with the output of the VCO being fed back to frequency control circuit 120 to be compared with F.sub.REF in the phase detector. The output of the phase detector is low-pass-filtered and then provided to VCO 110 and tuned filter 100.
Q-control circuit 130 is provided with a reference voltage V.sub.REF and typically includes a rectifier, an adder, and a low-pass filter. When Q-control circuit 130 is combined in a loop with VCO 110, a loss control loop (or sometimes an amplitude-locked loop) results, with the output of Q-control circuit 130 being fed back to VCO 110 as control voltage V.sub.CON. In theory, the two feedback loops in FIG. 1 may interact. However, if one loop is made much slower than the other one, the two loops can be considered decoupled. In order for the frequency control loop (phase-locked loop) to work effectively, the amplitude of the VCO output should neither be too small for the loop to detect zero-crossings, nor too large for a negative conductance that is part of the VCO to work in the linear region. Strong amplitude regulation may be realized by the loss control loop which is set faster than that of the frequency tuning loop. Thus, if a variable capacitor is used for frequency tuning, when the frequency tuning loop is much slower than the loss control loop, the capacitor can be considered invariant when analyzing the loss control loop.
In line with these assumptions, a VCO as depicted in FIG. 2A is chosen. FIG. 2A is a block diagram of a typical prior art loss control loop which includes VCO 210, rectifier 220, adder 230, and a circuit 240 having transfer function H(s). VCO 210 is depicted as an ideal fixed LC-tank circuit in parallel with G.sub.L, a conductance which models the loss of the tank at the oscillation frequency, and tunable negative conductance -G.sub.N. G.sub.L is the narrow-band equivalent loss contributed by inductor L and capacitor C. G.sub.N is the absolute value of the negative conductance which is controlled by V.sub.CON to tune out the loss. G.sub.N is assumed to increase monotonically with respect to V.sub.CON. The output of VCO 210 is provided to rectifier 220. The rectified output, V.sub.RECT, is subtracted from V.sub.REF in adder 230, and the difference is provided to circuit 240, a filter which provides gain and generates control voltage V.sub.CON to minimize the difference between the envelope of the VCO output, V.sub.ENV, and V.sub.REF. The envelope voltage, V.sub.ENV, is the low frequency component of the rectified output, V.sub.RECT. When V.sub.ENV is greater than V.sub.REF, G.sub.N is made smaller to make the tank lossy and reduce the envelope voltage, V.sub.ENV. If V.sub.ENV is less than V.sub.REF, G.sub.N is increased, thus making Q negative and increasing the envelope voltage, V.sub.ENV.
FIG. 2B illustrates a possible breadboard circuit realization of the prior art circuit of FIG. 2A. Circuit 240 is chosen as an integrator 270. (The signs on the inputs to adder 230 are reversed because of the minus sign introduced by integrator 270.) VCO 250 is realized as a differential LC-tank circuit in parallel with negative conductance circuit 260 which uses two MOSFETs biased in the triode region to act as variable resistors. Inserting resistors R.sub.1 and R.sub.2 in series with inductor L.sub.V reduces the Q of VCO 250 to about 10. Such a low Q is used to demonstrate the capability of the loop to control loss. Because this circuit is realized on a breadboard, the oscillation frequency of the tank circuit is low, around 3 MHz.
The goal of a loss control loop, when the loop settles, is to make Q infinite and the envelope voltage close to V.sub.REF. However, the loops of FIGS. 2A and 2B, which are adequate for kHz-range systems, have stability problems at higher frequencies. When the bandwidth of circuit 240 having transfer function H(s) is much smaller than ##EQU1## where ##EQU2## and ##EQU3## the control loop is unstable. Thus, as Q decreases or .omega..sub.o increases, the possibility of instability increases. For a low-Q tank circuit oscillating in the GHz range, a wide-bandwidth control required for the stability of the circuit is difficult to realize, and is not desirable due to the large high frequency leakage to V.sub.CON. Realizing H(s) as an integrator, as in FIG. 2B, is supposed to provide very high DC gain to make V.sub.ENV =V.sub.REF. However, non-linear analysis demonstrates that the loss control loop will settle at V.sub.ENV =V.sub.REF and G.sub.N (V.sub.CON)=G.sub.L only if these are also the initial conditions. If they are not the initial conditions, the loss control loop will be unstable. During part of the period, V.sub.ENV may be large enough to drive the negative conductance into its non-linear region, and during part of the period it may be too small to be detected by the frequency tuning loop. The results of such a loss control loop, as implemented in the breadboard circuit, are shown in FIG. 2C. The top trace shows VCO output voltage as a function of time and the bottom trace shows VCO control voltage V.sub.CON, as a function of time. In this instance, the peak-to-peak voltage of V.sub.CON is about 22 mV. Even though V.sub.REF is set to .about.0.1V, the peak-to-peak oscillation amplitude of the VCO rises to 1V, and is limited only because of the non-linearities of negative conductance circuit 260. Increasing the time constant of the integrator does not reduce this problem. Moreover, the loop is also unstable when the integrator is replaced by a low-pass filter with narrow bandwidth.
As realized on a breadboard, the prior art control method can not provide effective amplitude regulation in the MHz range or at higher frequencies. The amplitude of the VCO output may sometimes be too small for the frequency tuning loop to detect zero-crossings of the VCO's output, or too large for the negative conductances to operate in the linear region, causing frequency tuning mismatch between VCO 110 and tuned filter 100. It is expected that the same is true at microwave frequencies, as well. Therefore, a need exists for an improved loss control loop circuit which tunes the loss of the VCO precisely and achieves robust amplitude regulation.